DC-to-DC switching converters are widely used in the power electronics industry to serve as various AC-to-DC preregulators and DC power sources 1–4. Among many nonisolated topologies, the buck-boost converter is one capable of supplying an output voltage that can be stepped down or up from the input voltage 5–11. However, the voltage gain of the buck-boost converter is not a linear function of the duty cycle, and thus the control signal. Moreover, the small-signal control-to-output transfer function varies with the steady-state duty cycle. This is an undesirable feature when designing the feedback loop, especially if an adjustable output voltage is required. Computer simulations are performed to evaluate an adaptive feedback controller presented in 11. A nonlinear PWM control that requires the generation of an exponential carrier is proposed in 12 to improve the closed-loop performance. The control circuitry is somewhat complicated and the nonlinear modulating approach is an approximated solution rather than an exact one.

In this letter, the DC voltage gain and small-signal characteristics of a PWM buck-boost converter under continuous conduction mode (CCM) are reviewed. An indirect voltage command is introduced to exhibit a linearized DC voltage gain, and to eliminate the dependence of the DC gain of the control-to-output transfer function on the duty cycle 13. Experimental results on a prototype converter are given to verify the effectiveness of the presented control scheme.

2. PROPOSED INDIRECT CONTROL VARIABLE

Figure 1 shows a simplified schematic of the buck-boost power stage. R_{C} is the equivalent series resistance (ESR) of the capacitor. R represents the load seen by the power stage output. When the switch is turned on, the inductor is charged by the input voltage. When the switch is turned off, the diode provides a conducting path for the inductor current, and the inductor discharges to the load. Not considering the parasitics, the voltage gain under CCM is expressed as

(1)

where d is the duty ratio of the switch. Typically, the duty ratio is proportional to a control voltage, v_{d}, which is compared with a triangular or a sawtooth-like high-frequency carrier, v_{t}, to produce the gating signals. From (1) it is evident that the voltage gain is a nonlinear function of d and thus v_{d}. Moreover, at a higher output voltage, a small perturbation in v_{d} will result in a much larger swing in v_{0}. The nonlinear transfer characteristic between v_{d} and the voltage gain can be revealed in the following expression:

(2)

where V_{T} is the peak value of the carrier signal v_{t}.

The small-signal control-to-output transfer function in the s-domain is 14

(3)

where and are the small-signal expressions for v_{0} and v_{d}. D is the steady-state value of the duty cycle. Q and ω_{0} are the quality factor and the natural frequency. ω_{z1} is a zero due to the ESR of the capacitor. ω_{z2} is a right-half-plane (RHP) zero which depends on the load level and the duty cycle.

(4)

(5)

(6)

(7)

It is clearly seen that a term relating to the duty ratio is included in the denominator in (3). Therefore, the DC gain of T(s) is varied as the duty ratio changes, which is unfavorable for the feedback loop design with an adjustable output voltage.

From (2), to achieve the linearity between a new control signal v_{d}′ and the voltage gain, it is required that

That is, the new control voltage or the indirect control voltage, v_{d}′, is first multiplied by the peak of the carrier signal, which is usually a constant, then divided by the sum of V_{T} and v_{d}′ itself. The result is fed into the PWM comparator. The block diagram illustrating the proposed control mechanism is shown in Figure 2. From (8), the small-signal ratio between these two control variables can be derived. It is assumed that

(10)

(11)

where variables in upper-case represent the DC quantities and the italic lower cases with a cap are the small signals. Substituting (10) and (11) into (8), and neglecting the product of two small perturbations yields

By substituting (14) into (3), the modified small-signal control-to-output transfer characteristic with the new control voltage v_{d}′ becomes

(15)

From (15), it is noticed that for a constant input voltage, as V_{T} is constant, the incremental DC gain remains constant for the CCM buck-boost converter under the proposed control method. Also, the locations of the dominant open-loop zeros and poles are theoretically unchanged. There is no need to reconsider the issues of the stability and dynamic response.

3. EXPERIMENTAL RESULTS

A prototype converter is built to perform the experiments with the following parameters: V_{i} = 50V, L = 2.85mH, C = 300µF/450V, and R = 100Ω. The switching frequency is 60 kHz. One possible implementation of the block diagram in Figure 2 is shown in Figure 3. TL494 is adopted as the PWM comparator. V_{T} is about 3.35 V in a TL494. A low-cost analog multiplier, such as AD633, is placed in the feedback loop of a noninverting amplifier to perform the division operation in (9) 15. The output of the multiplier is obtained as

(16)

As the multiplier is in the feedback path of the noninverting amplifier, W can also be derived as

By setting R_{1} = R_{2} = R_{f} = 10kΩ and R_{3} = 49.7kΩ, M is calculated to be 3.35, which is equal to V_{T}. Therefore, the conversion between v_{d} and v_{d}′ as described in (9) can be fulfilled.

Figures 4(a) and (b) show the recorded nonlinear and linear transfer characteristics between V_{0} and the control voltages. A fairly constant gain is observed for the whole operating range in Figure 4(b).

Figures 5(a) and (b) depict, respectively, the open-loop frequency responses (the gains are scaled down by 42, or equivalently 32 dB, for measurements) under the conventional and the proposed control schemes for V_{i} = 120V, D = 0.4, and V_{0} = 80V. The dominant pole frequencies (at the phase of − 90^{∘}) for both cases are around 100 Hz. It is thus concluded that the transients of the compensated system are stable as well. The DC gain for the proposed system reduces slightly from 8 to − 0.9dB. The difference of the DC gain, − 8.9dB, is equivalent to (1 − D)^{2}.

Figure 6 shows the output voltage regulation under a step load current change with a proportional (P) feedback loop. In Figure 6(a), when I_{0} drops suddenly from 2 to 0 A with the conventional control method, v_{0} rises and then stabilizes within 50 ms. The same dynamic performance can be observed in Figure 6(b) for the proposed system. However, the proposed system has a larger steady-state error due to a smaller DC gain. Nevertheless, this nonzero steady-state error can be eliminated by applying a proportional-integral (PI) feedback technique.

4. CONCLUSION

A novel method for improving the control-to-output transfer function of a PWM buck-boost converter is realized in this letter. An indirect control signal is introduced. From the DC and the small-signal characteristics of the discussed buck-boost converter, the linearization of the steady-state voltage gain is achieved. Also the dependence of the DC gain of the control-to-output transfer function on the duty cycle is eliminated. The implementation of the presented controller requires additional analog IC's. Nevertheless, the proposed control method can be implemented via digital approach as well, and applied to any buck-boost derived converter topologies.