The impact of an azimuthal slot in the antenna flair on the performance of a bowtie antenna has been analyzed. It has been shown that the slot is equivalent to resistance-capacitance loading. The resistance of the slot is physically caused by radiation of electromagnetic energy from the slot. Equivalent resistance and capacitance of the slot have been analyzed against geometrical parameters and frequency. The azimuthal slot has been used to improve the bowtie performance and to optimize this antenna for an ultrawideband impulse radio system under development. After optimization, the antenna exhibits a better performance than the conventional bowtie in terms of efficiency, late-time ringing, and matching to the generator/receiver. The performance of the optimized antenna has been verified experimentally; good agreement between theory and experiment has been observed.
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 Ultrawideband (UWB) communication is a promising technology that is expected to solve the problem of the shortage of the available frequency bands in wireless, short-distance networks. It is characterized by a potentially high capacity of the channel, low spectral density of the transmitted power (and thus low energy consumption), and high immunity against electromagnetic (EM) interference and fading robustness [Roy et al., 2004]. One of the realizations of UWB communication is the so-called impulse radio, in which, similar to optical communications, information is transmitted by means of physical radiation of very short EM pulses (of the order of some hundreds of picoseconds).
 It is expected that future short-range indoor UWB telecommunication systems will operate in the frequency band from 3.1 to 10.6 GHz according to the Federal Communications Commission (FCC) mask [Federal Communications Commission, 2002]. Antennas for such a system should have an operational band of at least a few gigahertz within the above-mentioned frequency band. Moreover, to radiate short pulses without substantial late-time ringing (oscillations in the radiated waveform after the main pulse), antennas for the impulse radio should have a linear phase characteristic within this frequency band. From a possible application point of view, almost omnidirectional radiation patterns, at least in a half-space, for such an antenna are desired. And to reduce power consumption of the whole system, antenna efficiency should be high. For the impulse radio, the antenna efficiency is determined by a peak-to-peak magnitude of the radiated pulse (and not by a total radiated energy), as the performance of the majority of time domain receivers depends on the ratio of the pulse peak amplitude to the root-mean-square value of noise.
 Furthermore, there are several technological demands on the antennas for indoor UWB telecommunication. First, the antenna should be small and flat (two-dimensional) to be easily integrated into a small (mobile) device. Second, the antenna should be closely integrated with a transmitter and a receiver (antenna on a chip is an ideal option). Thus the antenna should be very well matched to the generator and the receiver circuit (VSWR should be less than 2 within the whole operational band). Integration with RF circuits gives, however, additional freedom in antenna design, as the antenna input impedance should not be limited to 50 ohms, and balanced antenna feeding can be realized without a balun by using a differential amplifier in the receiver and by using a generator with a differential output in the transmitter. Finally, the antenna should be mounted on a dielectric substrate, which will also serve as a protective mechanical shield, and should radiate through it.
 There are no available antenna designs which satisfy all above-mentioned parameters. However, optimization of known types of transient antennas can lead to success [Schantz, 2003; Yarovoy et al., 2004]. The compromise between operational bandwidth and antenna efficiency, antenna physical dimensions, and ringing becomes the main research issue [Schantz, 2004].
 The most widely used technique for enlarging antenna bandwidth and for reducing late-time ringing is the application of resistive loading, in which the well-known Wu-King profile [Wu and King, 1965] can be used to determine the loading distribution along the antenna. Because of difficulties in the practical realization of nonlinear loading profiles with complex-valued loading (e.g., Wu-King) on a planar structure [Shlager et al., 1994], purely resistive loading is widely applied. The main disadvantage of resistive loading is that it reduces radiation efficiency considerably. In the case of the Wu-King profile, it has been shown that radiation efficiency decreases to a level as low as 23% [Maloney and Smith, 1993]. To avoid this drawback, the use of nondissipative reactive loading has been proposed. Nyquist and Chen  suggested inserting a pair of lumped reactive elements in order to enlarge antenna bandwidth around a certain frequency by placing the elements at an optimal distance from the antenna ends. However, it has been shown that resistive loading is the most efficient one for suppressing late-time ringing, while capacitive loading (alone or in combination with resistive loading) still exhibits a relatively high level of ringing [Montoya and Smith, 1996].
 In this paper we investigate a loading technique, which substantially enlarges the antenna bandwidth and reduces antenna ringing while keeping antenna efficiency high. This technique ultimately suits two-dimensional antennas (all modifications of the bowtie antenna). The technique employs an azimuthal slot in the antenna flairs. In our earlier work [Yarovoy et al., 2002] we have introduced a straight slot in the antenna flairs to improve the bowtie efficiency, and we have combined azimuthal slots with a volumetric absorber in order to realize a distributed resistance-capacitance (RC) loading [Lestari et al., 2004]. In this paper we investigate in detail the impact of the azimuthal slot on surface currents and radiation from a bowtie antenna and optimize the antenna for impulse radio applications. A numerical model of the antenna is briefly presented in section 2. The impacts of the slot and the dielectric substrate on the antenna performance are presented in sections 3 and 4, respectively. The optimized design is described in section 5, while its experimental verification is presented in section 6. Discussion of the results achieved as well as conclusions are presented in section 7.
2. Numerical Model
 Let us consider a bowtie antenna with a flair angle of 90° and with azimuthal slots of width Δ in both flairs (Figure 1). The distance between the feeding point and the slot is l0, and the distance between the slot and the antenna termination (the strip width) is l1. To perform the above-mentioned analysis and optimization, we have developed a computational model using the commercial EM simulator FEKO (its description is available at http://www.feko.info), which is based on the mixed-potential integral equation formulation. Antenna flairs, which are assumed to be perfectly conducting and infinitely thin, are subdivided into triangular surface elements. We have tested two feeding models: the delta gap model and the edge feed model [Leat et al., 1998]. We did not find any substantial differences between them, especially for the far-field values. For reasons of simplicity, we have selected the delta gap model. The impact of the infinite dielectric substrate is taken into account by the proper Green functions in the kernels of the integral equations. The integral equation is solved by the method of moments.
 From a system design point of view, the antenna should be integrated with a pulse generator on a chip, which should be placed between antenna flairs. That is why the feeding line is not included in the numerical model. We assume that the generator fires a pulse, whose waveform can be described as the first derivative of a Gaussian pulse with the duration of 0.2 ns defined at a level of 10% of maximal amplitude. The spectral content of the pulse is already insignificant (slightly higher than −40 dB with respect to the maximum) at 20 GHz. Hence simulations have been done in the frequency domain over 101 frequencies from 200 MHz to 20 GHz. To improve the time domain resolution, we applied zero padding, expanding the frequency range up to 49.4 GHz. Together with a proper windowing in the frequency domain and the inverse Fourier transform for calculating the radiated field in the time domain, this approach allows us to perform a fast and accurate time domain analysis of antennas with arbitrary-shaped, metal flairs embedded into a multilayer dielectric structure.
3. Azimuthal Slot as Loading Element
 In this section we will investigate the impact of an azimuthal slot on the surface current distribution and will demonstrate that the slot plays a role of complex load. The current on antenna flairs mainly flows in azimuthal direction from the feeding point to the flair termination. A slot in the flair introduces a discontinuity in the antenna and distorts the electromagnetic field around it. Let us call this distortion scattered field ESC. According to simulations, the scattered field is concentrated around the slot (Figure 2), and this concentration becomes more pronounced with frequency increase. Numerical analysis also shows that the electric field is reasonably constant through the slot if Δ ≪ λ (Figure 3a). Frequency increase causes both increase of the absolute value of the scattered field in the slot and inhomogeneity of the field distribution over the slot (Figure 3b).
 There is no surface current in the slot. However, there is a displacement current , which is equal to the time derivative of the electric displacement in the slot [Collin, 1991]. The total flux of the displacement current through the slot can be found by the following expression:
where Eρ is the radial component of the scattered field in the cylindrical coordinates introduced in Figure 1, and integration on azimuthal angle ϕ is performed over all angles Φ corresponding to the antenna flair. In equation (1) we assume that the scattered field at ϕ ∉ Φ is negligible. Assuming further a cylindrical symmetry for the scattered field at ϕ ∈ Φ and time dependence exp (jωt), one can derive from equation (1)
On the other hand, the displacement current can be written as
where Y is the complex-valued (Y = Y′ + jY″) admittance of the slot, and U is the voltage across the slot
Comparing equations (2) and (3), one can conclude that the slot has the same impact on the surface current as RC loading (parallel connection of resistor and capacitor), where the slot equivalent resistance R ≡ 1/Y′ equals
The resistance of the slot is physically caused by radiation of electromagnetic energy from the slot. The imaginary part of the slot admittance can be represented as Y″ = ωC, where the equivalent capacitance of the slot is given by
Variation of the electric field over the slot causes minor influence (less than 10%) on the values of the equivalent parameters of the slot. For example, for a frequency of 5 GHz and values of geometrical parameters l0 = 8 mm, l1 = 10 mm, and Δ = 1 mm, by using equations (5) and (6), we obtain R = 416.63 ohms and C = 0.4048 pF. If we neglect a variation of the electric field in the slot by using instead of equation (4) an approximate expression for the voltage across the slot U,
we arrive at the approximate values of the slot equivalent resistance and equivalent capacitance R = 383.5 ohms and C = 0.450 pF. Since the approximate values differ less than 10% from the values calculated by using equations (5) and (6), we shall use the approximation (7) for computing the voltage across the slot below.
 The dependence of the equivalent parameters of the slot (with a width of 1.1 mm) on the frequency is shown in Figure 4a. The equivalent resistance decreases with the frequency. Within the frequency band from 3 to 6 GHz, the resistance is inversely proportional to frequency squared. Taking into account that for the above-mentioned frequencies the slot width remains small compared to the wavelength, the observed frequency dependency can be explained by the linear increase of both the total electrical length and the electrical width of the slot with frequency. The equivalent capacitance remains almost constant over the frequency band because the increase of the capacitance due to the linear increase of the electrical length of the slot with frequency is compensated by the decrease of the capacitance due to the linear increase of the electrical width of the slot.
 The equivalent resistance of the slot linearly increases with its width (Figure 4b). The equivalent capacitance decreases while the slot gets wider (similar to conventional capacitors). The equivalent resistance of the slot depends only slightly on the distance l0 between the feeding point and the slot, while the equivalent capacitance linearly increases with l0. The latter phenomenon reflects a decrease of the slot curvature and an increase of the slot length. Finally, the equivalent capacitance is almost independent of the strip width l1.
 The impact of the azimuthal slot on the input antenna impedance is shown in Figure 5. It can be seen that the slot substantially flattens both the real and imaginary parts of the input impedance and transforms them into monotonically increasing functions of the frequency. Furthermore, in the frequency band from 3 to 9.5 GHz, the slot decreases the input impedance of the antenna, which results in better matching of the antenna to a pulse generator.
4. Impact of Dielectric Substrate
 So far, we have discussed the behavior of an RC-loaded bowtie antenna in free space. However, the technological demands require that the antenna should be mounted on a dielectric substrate. To understand the impact of the substrate on the antenna characteristics, we have calculated the input impedance and the antenna gain for different values of the dielectric constant and the substrate thickness. The values of the substrate parameters have been chosen to match the commonly used Duroid 5870 and Roger 4003 substrates. Substrate size along the antenna has been assumed to be infinite. Antennas on finite substrate have been analyzed by Vorobyov et al. , who showed that there is almost no difference between finite and infinite substrates if the substrate size is at least 1.5 times larger than the flair size. The antenna dimensions have been assumed to be l0 = 8 mm, Δ = 0.2 mm, and l1 = 10 mm.
 If we consider a constant thickness of the substrate, an increase of its dielectric permittivity ɛr causes a shift of the input impedance curves toward lower frequencies (Figure 6a). The shift can be explained by two physical phenomena. First, the presence of a dielectric substrate causes redistribution of the surface current: The current on the surface facing the dielectric is larger than the current on the opposite surface. Second, the propagation velocity of the surface current on the dielectric side is slower. Thus, in terms of electromagnetic radiation, the antenna becomes electrically larger. In addition to the shift in frequency, we observe a decrease of the input impedance value, both the real and imaginary parts, by increasing the value of ɛr.
 Increase of the substrate thickness at a fixed value of its dielectric permittivity does not have an essential impact on the input impedance for the frequencies below 6 GHz (Figure 6b). For higher frequencies, an increase of the substrate thickness is similar to an increase of the dielectric permittivity and causes a shift of the input impedance curves toward lower frequencies.
 Regarding antenna radiation, an increase of the substrate dielectric permittivity ɛr, by a constant thickness of the substrate, slightly decreases the antenna gain and shifts a dip in the gain curve toward lower frequencies (Figure 7a). The latter is caused by an increase of the antenna electrical length with an increase of substrate dielectric permittivity and lowering the frequency, at which the antenna pattern splits into two lobes. An increase of substrate thickness (by a constant dielectric permittivity) also shifts the gain dip toward lower frequencies (Figure 7b).
5. Optimization of Antenna Parameters
 The optimization of the antenna design described above has been done via numerical simulations. The optimization criteria were amplitude of the radiated pulse by a given output from a pulse generator (which for transient antennas determines the efficiency of antennas and depends on the matching to the generator) and the late-time ringing. The former we characterize by the peak-to-peak amplitude of the radiated pulse at the distance 1 m away from the antenna. To characterize the latter, we introduce a new parameter called the pulse extension ratio (PER). We define it as
where Trad is the time interval within which the magnitude of the pulse radiated in the boresight direction decreases by 40 dB, and Tgen is the duration of the output pulse from the generator defined at a level of 10% of maximal amplitude. From the system design point of view, an acceptable level of the antenna ringing is about –40 dB with respect to the main pulse amplitude after triple duration of the pulse. Therefore the desired value for the PER is about 3.
 With respect to optimization our goal was to increase the magnitude of the radiated pulse (keeping close to omnidirectional radiation patterns) and to decrease the PER. In order to achieve this, we varied the antenna's dimensions and the type of the substrate. We found that for the antenna placed on a substrate with dielectric permittivity of 3.4 and thickness of 0.8 mm, a distance of 0.8 cm from the feeding point to the slot leads to the best result in terms of magnitude of the radiated pulse. In this case, the radiation from the slot constructively adds up with the radiation from the feeding point. This result agrees well with Lestari et al. , who found that the optimal distance to the first slot should be equal to a quarter of the wavelength at the pulse central frequency. In our particular case, because of the presence of the substrate, the electromagnetic field propagates along the flair with a velocity lower than the speed of light (see section 4), and thus the slot should be closer to the feeding point. The combination of the slot width of about 0.2 mm and the strip width of about 1 cm leads to the best compromise of low ringing and high efficiency. The input impedance of the antenna with such a slot well matches the output impedance of the pulse generator. As a result, the magnitude of the radiated pulse is larger than that for a solid bowtie. With respect to the substrate, we have found that the Roger 4003 substrate with dielectric permittivity of 3.4 and thickness of 0.8 mm provides the desired radiation patterns and an almost constant value (of about 150 ohms) of the input impedance in the desired operational frequency band. The manufacturing of a pulse generator on a chip with differential output and differential output impedance of 150 ohms does not create any extra problems in comparison with the standard 50 ohm output impedance case. Besides, because of the integration of the antenna and a generator, the allowed values of the antenna input impedance are not limited to the values of the characteristic impedances of the available transmission lines.
 A simulated performance of the optimized antenna is presented in Figure 8. The magnitude of the radiated pulse is almost 1.37 times larger than the one radiated by a conventional bowtie antenna of the same size. The PER parameter is about 7.5, which is 1.57 times smaller than for a conventional bowtie of the same size (PER = 11.75). The waveform of the radiated pulse in the far field remains almost the same in both E and H planes for all elevation angles (Figure 9). The magnitude of the radiated pulse remains almost constant in the H plane for all elevation angles and gradually decreases with the elevation angle in the E plane. The radiation patterns (as defined in the frequency domain) of the antenna are close to those of a resistively loaded dipole within the desired frequency range (Figure 10).
6. Experimental Verification
 The theoretical design has been verified by the following measurements. To avoid problems with antenna miniaturization, the antenna has been manufactured on a 0.8 mm thick Roger 4003 substrate with 4.5 times larger dimensions than mentioned above. The substrate size is about 13 cm by 3 cm. The antenna is fed by a double semirigid cable (each cable has 50 ohm impedance) (Figure 11). Such antenna feeding closely resembles the expected differential feeding of the actual antenna and avoids the use of a UWB balun. To support the feeding cable and to mount the antenna in the anechoic chamber, a PVC support has been introduced.
 The antenna input impedance is shown in Figure 12a, while the antenna gain is shown in Figure 12b. In general, a good agreement between simulated (for the enlarged version of the antenna) and measured parameters can be observed. Some discrepancies are caused by the common mode current, which flows on the feeding cables. Reflection of this current from the antenna support is responsible for oscillations of the measured S parameters and the antenna gain around theoretically predicted values.
7. Discussion and Conclusions
 We have optimized a bowtie antenna to satisfy the demands of the UWB impulse radio systems. In order to improve the transient performance of the bowtie, we have used an azimuthal slot in the antenna flairs. Similar to the resistive loading, the slot enlarges the operational bandwidth of the antenna. Contrary to the resistive loading, the slot does not decrease the antenna efficiency as defined in the time or frequency domains because it does not absorb any energy of the radiated field.
 We have analyzed the impact of an azimuthal slot on the antenna performance. We demonstrated that the slot is equivalent to the loading of the antenna by the resistor and capacitance in parallel. The resistance of the slot is physically caused by radiation of electromagnetic energy from the slot. Equivalent resistance and capacitance have been analyzed against geometrical parameters and frequency. The proposed approach can be used for optimization of multislot antennas [Lestari et al., 2004]. To this end, it is important that equivalent parameters of the slot hardly depend on the strip width l1.
 The results of numerical optimization are encouraging. A good matching of the antenna to a 150 ohm generator has been achieved in the frequency band from 6 to 10 GHz, which is the upper part of the operational band specified by the FCC for indoor UWB communications. Because of the better matching to a pulse generator, the antenna efficiency (in the time domain) is 1.37 times larger than that of a conventional bowtie antenna of the same size. The duration of the radiated signal (at the level –40 dB) is 1.57 times smaller than by a conventional bowtie. The performance of the optimized antenna has been verified experimentally.
 The authors are grateful to P. Hakkaart for the manufacturing of the scaled version of the antenna and to J. H. Zijderveld for his help with the antenna measurements. This research is supported by the Technology Foundation STW, the applied science division of NOW, and the technology program of the Ministry of Economic Affairs of the Netherlands.