Magneto‐Mechano‐Electric Antenna for Portable VLF Transmission

Wireless communication has always been an indispensable element in the modern information‐based society. Beyond the commercial electrical antenna, very low frequency (VLF) mechanical antennas have recently become research hotspot since their combination of miniaturization and favorable radiation efficiency in lossy electrically conductive environments. However, their usage is challenged due to the relatively limited radiation capability and modulation bandwidth. This study demonstrates an improved high‐efficiency magnetoelectric (ME) mechanical antenna based on the magneto‐mechano‐electric (MME) effect, realized via the synergistic effect of piezo‐driven magnets motion and converse magnetoelectricity (ME). Converse ME coefficient and radiation measurement show that the MME antenna demonstrates superior performance over a normal ME antenna. The oscillation magnet serves as an extra vibrating magnetic dipole besides the ME composite and thus brings radiation enhancement to the mechanical antenna. Furthermore, digital signal modulations are conducted with a VLF carrier signal to enable anti‐interference and anti‐attenuation communication. In view of the exemplary demonstration, the MME effect‐based antenna is expected to provide a new strategy for mechanical antenna improvement and shows tremendous potential for conductive environments communication applications.


Introduction
For the last hundred years, humanity's incremental understanding of electromagnetic fields and electromagnetic waves has resulted in vigorous technological development resulting in modern antenna technology. The wide application of wireless communication technologies has enabled ubiquitous application spaces including smart phones, broadcasting stations, radio frequency identification systems, and radars. [1,2] As one of the omnipresent critical components in the radio frequency system, antennas have long been the focus of research and design. Current state-ofthe-art systems have successfully realized high quality wireless communication in atmospheric environments relying on antenna technology. Applications in emerging environments, such as underwater, underground, inside the human body, or in other lossy, electrically conductive environments put forward new demands for antenna transmission capability that cannot be met using traditional antenna technologies. [3,4] When electromagnetic waves propagate in these conducting medium, the dielectric loss increases with frequency, and the electromagnetic waves are attenuated, leading to significant reduction in effective communication distance. [5] To mitigate highfrequency loss mechanisms, very low frequencies (VLF) operations (3-30 kHz) have been proposed as a potential solution due to the increased signal penetration depth in these conducting mediums. [6][7][8] Current conventional antennas are directly driven by electric current or voltage to accelerate the electrons inside the metal plates for radiation. [1] Conventional electric antennas rely on electromagnetic resonance for efficient operation, and are typically designed to be larger than one-tenth of the electromagnetic (EM) wave wavelength. [9][10][11] Such a limitation requires antenna towers over 100 m to achieve 1-10 km wavelengths of VLF transmission. [12] To enable practical VLF communication, it is imperative to develop technologies that can enable VLF antenna system miniaturization and portability. [4,5] Although electrically small antennas do show some advantages due to their compact size and low cost, the relatively low www.advancedsciencenews.com www.advelectronicmat.de radiation efficiency and low bandwidth still remain as obstacles for practical applications in VLF communication. [2,13,14] In contrast to the most widely used dipole antenna, mechanical antennas that function through the coupling of the electromagnetic field by using the mechanical movement of charge or magnetic dipole have gained prominence. [3,[15][16][17][18] One approach to achieve a mechanical antenna is to physically rotate or vibrate a permanent electret. It is worth noting that antennas based on the motor driven electret exhibits favorable extremely low frequency (0-200 Hz) band radioactive performance with significantly reduce of the size. [19] Similarly, mechanical magnetic antennas have been developed by utilizing a spinning magnet. [4,15,20] In the near field of magnetic antenna, the electromagnetic radiation is predominantly comprised of magnetic fields rather than electric fields, which shows substantially smaller near-field losses and propagate signals to larger distances with losses up to three orders of magnitude smaller in lossy electrically conductive environments. Previous studies have confirmed ≈1 pT magnetic radiation at the frequency of 100-500 Hz can be observed at a distance over 100 m from magnetic antenna. [15] However, both of the physical electret and magnet based mechanical antennas require motors to provide large inertial forces that make signal modulation challenging and result in low bandwidth. [3,17,21] Under these circumstances, Prasad et al. proposed a portable electromechanical system called a Magnetic Pendulum Array. The system eliminates the motors by utilizing a coil to drive an array of magnetic pendulums in oscillatory motion. This approach allows the system to achieve a significantly higher quality factor than conventional coils and thus obtain order of magnitude higher transmission efficiency, which demonstrates attractive application potential in underwater communications and underground localization. [22] It is worth noting that Mechanical antennas can also be excited by self-resonant mechanical structures based on piezoelectricity or magnetoelectricity, [23][24][25][26] which rely on the oscillating electric/magnetic dipoles serving as radiation source. A VLF transmitter based on a high Q lithium niobate piezoelectric crystal rod has been introduced with the Direct Antenna Modulation method, which has shown higher radiation efficiency than comparable conventional antennas. [10] Moreover, using high dielectric constant PZT piezoelectric ceramics is found to increase the electric dipole density and leads to enhancement of radiation efficiency. [23] The high Q-factor of these devices resulted in antennas that are limited due to an efficiency-bandwidth tradeoff. On the other hand, multiferroic magnetoelectric (ME) composites demonstrate fascinating phenomenon in electricmagnetoelectric conversion, promising application in various functional devices, including antennas. [27][28][29][30][31][32] Schneider et al. report a multiferroic antenna using a PZT stack to produce dynamic axial compressive stresses in a FeGa rod. [33] Similarly, Sun et al. designed a bulk magnetoelectric (ME) antenna composed of PZT fibers and Metglas (FeSiB) films. [10] The nonlinear antenna modulation method improves the transmission data rate and decouples the information bandwidth and antenna efficiency for the ME antenna. [34] Compared with permanent magnet or electret based mechanical antenna, these acoustically driven mechanical antennas demonstrate advantages in miniaturization and signal modulation, though the radiation capability is relatively weak.
In this work, a VLF transmitter based on the magnetomechano-electric (MME) effect has been designed and demonstrated. The system consists of a fixed PZT/Metglas stack with permanent magnet attached to its free end. Measurement of ME coupling properties and electromagnetic radiation properties have been conducted, followed by analog and digital signal transmission tests. Experimental results indicate that the tip magnet not only provides magnetic bias for the ME composite, but also creates significant electromagnetic radiation gain due to its vibration. Such enhancement in radiation caused by MME effect is expected to provide promising strategy for ME antenna improvement.

Working Principle and Design of Magneto-Mechano-Electric (MME) Antenna
To obtain maximum radiated signal efficiency, the acoustically actuated antennas are composed of a high Q-factor resonator, in which the extremely low mounting losses, external dampening, and internal losses in the piezoelectric material itself together will significantly improve the performance of the antenna. Under this consideration, PZT-4 ceramic with a Q-factor of ≈1000 is chosen as the piezoelectric phase. Longitudinally magnetized Metglas films are attached on the two sides of PZT to make up a sandwich structure ME composite, as shown in Figure 1a. In order to construct magneto-mechano-electric (MME) structure, the as made ME composite beam is designed to become cantilever structure with a fixed boundary. As illustrated in Figure 1b, the multiferroic ME composite demonstrates electrically controlled magnetic domain wall motion using strain-mediated magnetoelectric coupling in piezoelectric/ferromagnetic structures. The magnetic domain motion results in local magnetization change, as would magnetic dipole vibration. Clearly, increasing the density and oscillation amplitude of the magnetic dipoles can effectively enhance the radiation intensity. Here we attach a permanent NdFeB magnet to the free end of the ME cantilever ( Figure 1a). When driven by the piezoelectric phase of the laminate, the generated magnetic moment m 0 will serve as extra radiation source to increase the overall radiation intensity of the antenna. Compared to other permanent magnet based mechanical antennas, the entire MME antenna system shows advantage in compact size, which makes it more suitable for portable communication.
Finite element analysis (FEA) of induced magnetic flux density distribution has been conducted to verify the enhancement of the system due to the influence of the permanent magnet. It can be seen that the fixed MME cantilever (Figure 1c-iii) produces much stronger spatial magnetization distribution compared to stress-free L-T sample and fixed ME cantilever (Figure 1c-i,ii). Therefore, the MME effect introduced by permanent magnet provides prospect in optimization of high efficiency ME antenna. To evaluate the signal transmission capability, a whole communication link is built and shown in Figure 1d,e. With the measurement setup, it is significative to evaluate the potential of using MME antennas in practical wireless signal transmission application. Design and working principle of the very low frequency (VLF) magnetoelectric (ME) antenna. a) Schematics of the ME composite as a VLF antenna. b) Schematic representation of the electromagnetic radiation excited by the magneto-mechano-electric (MME) effect induced by ME composite and oscillating magnet. c) Finite element analysis of induced magnetic flux density vectors. i) Stress-free L-T mode ME laminate, ii) one-end clamped ME cantilever and iii) one-end clamped MME cantilever d) Electrical schematic and e) setup of magnetoelectric VLF communication system.

ME Performance of PZT/Metglas Composite and MME Antenna
Since the antenna radiation is directly related to strain of the magnetostrictive phase, we characterized the voltage-induced strain of the ME composite. Figures S2 and S3 (Supporting Information) show both FEA and experimental results of DC electric fieldinduced strain from the pristine PZT-4 ceramic and the L-T mode ME composite with different numbers of Metglas films (up to 20 pairs of Metglas layers). The strain data reported in Figure S3 (Supporting Information) were measured using a microstrain gauge attached to the surface of the PZT and Metglas layers, respectively. Considering most of the antenna tests are conducted in air environment, the test voltages are limited to 1.1 kV to avoid electric breakdown. The strain is decreased with increasing num-ber of Metglas layers due to increased stress loss from incremental epoxy resin and Metglas layers. Strain induced by the PZT beam is still apparent at the surface of ME composite, even when there are 20 layers of Metglas in the structure. None of the strains for the pure PZT or Metglas layers reach to their saturated values at a DC voltage of 1.1 kV or less, which means the ME composite can likely still generate magnetic domain motion beyond the 1.1 kV voltage range tested here.
The direct ME (DME) coupling and converse ME (CME) coupling properties of L-T mode ME composite were measured and are presented in Figure S3c,d (Supporting Information). Both the DME coefficient and CME coefficient increase with increased number of Metglas layers for a fixed PZT thickness. However, overmuch Metglas layers can lead to impaired CME coupling, and the increasing epoxy resin adhesive brings high mechanical loss. [35] The ME composite developed in this work therefore consists of 20 layers of Metglas layers and PZT-4 ceramic beam.
To maximize ME coefficient and thus radiative efficiency of the MME structure, it is essential that the attached magnets provide optimal magnetic bias. [31,36] The optimal magnetic bias field for the MME structure was measured to be ≈38 Oe ( Figure S6, Supporting Information), as tested by measurement system demonstrated in Figures S9 and S8 (Supporting Information). To determine the optimal size of the bias magnets, the DC magnetic field distributions in of the magnetostrictive phase generated by various tip magnetic were investigated using FEA (Figures S7 and S8, Supporting Information). We constrained our analysis by setting the size of magnet attached to the fixed boundary at 15 mm in diameter and 2 mm in thickness. We further constrained the diameter of the free end magnet to 8 mm and varied the thickness to determine optimal bias. As the thickness of the free end magnet increases, the magnetic field distribution in the magnetostrictive phase also increases. Our simulations suggest that a 2 mm thick free end magnet most closely approximates the optimum bias magnetic field of ≈38 Oe, the maximum quasi-static CME coefficient of the MME cantilever is nearly identical to that of the ME cantilever (0.17 Gs V −1 as can be seen in Figure 2a), thus we confirmed the optimal size of 8 mm in diameter, 2 mm in thickness, and 1.5 g in weight for the free end magnet.
The natural resonant frequency of the cantilever directly determines the central operating frequency of the antenna. The impedance dependence of the resonant frequency was modeled using FEA and measured empirically to quantify the electromechanical properties of composites with different structures (see Figure S4, Supporting Information). For the L-T mode ME composite, the electromechanical resonance frequency is 24.6 kHz, as shown by the phase inversion in Figure S4 (Supporting Information) and the ME peak in Figure S5 (Supporting Information). The electromechanical resonance frequency of ME cantilever is 11.5 kHz, significantly lower than the L-T mode.
The undamped natural frequency for a cantilever can be defined as: [36] = √ k eq where k eq is the equivalent spring rate, m eq is the equivalent mass, YI is the flexural rigidity of the beam, L is the length of the beam, m is the mass per unit length, and M t is the tip mass. The 33/140 term arises from the fact that only part of the beam mass participates in the motion. Note that here is the circular frequency, and the resonant frequency f = /2 . We therefore expect that the natural frequency of the cantilever system will be reduced by adding the bias magnets to the free end of the beam. Both FEA and experimental results ( Figure S4e,f, Supporting Information) show that the electromechanical frequency of MME cantilever decreases relative to the ME cantilever, i.e.,11.25 kHz versus 11.5 kHz, respectively. Converse ME coupling properties of the one-end clamped ME and MME cantilever. a) Quasi-static converse ME coefficient as a function of magnetic bias field. Inset is the magnetic induction of the one-end clamped MME cantilever generated in response to applied voltage. b) Dynamic converse ME coefficient as a function of frequency. Note that solid dots designate the experimental data, while the lines designate the theoretical data.
The converse ME coefficient of the MME cantilever structures can be calculated by: where mB 1 is the magnetic induction, E 3 is the electric field applied on the composite, q 11 is the pseudo-piezomagnetic coefficient of the magnetostrictive phase, t 1 is the thickness of the ME composite, m Y and a Y are the Young's modulus of the magnetostrictive phase and magnets, respectively, p 1 is the volume fraction of the piezoelectric phase, d 31 is the transverse piezoelectric coefficient, w is the width of the ME composite, L p is the length of the ME composite, p s 11 is the compliance coefficient of the piezoelectric phase, c 11 is the effective stiffness coefficient of the ME composite, D is the diameter of the free end magnet, 1 = L p k 1 , Figure 3. Near-field B-field patterns created by one-end clamped ME and MME cantilever. a) ME cantilever as a dipole in spherical coordinate system. b) Near-field radiation patterns measured on yz plane. c) Near-field radiation patterns measured on xz plane. d) Near-field radiation patterns measured on xy plane. e) MME cantilever as a dipole in spherical coordinate system. f) Near-field radiation patterns measured on yz plane. g) Near-field radiation patterns measured on xz plane. h) Near-field radiation patterns measured on xy plane. 1 and a are the density of the ME composite and magnet, respectively, and is the angular frequency. More details can be found in Section S1 (Supporting Information).
The experimental resonant ME coupling properties are shown in Figure 2b. Although slight difference between experimental and theoretical data is observed, which is probably caused by alumina plate fixed boundary and interfacial mechanical loss between different elements, while both experimental and theoretical results suggest a significant enhancement for the MME cantilever relative to ME cantilever. At the resonance frequency of 11.3 kHz, the resonant CME coupling for the MME cantilever is 46 Gs V −1 , ≈300% higher than the ME cantilever.
Based on the above results, the strength of magnets to ME antenna is reflected in the following aspects: i) The magnets provide bias magnetic field for the ME composite, ensuring that the magnetostrictive phase can produce maximum radiation driven by the electric field-induced strain. ii)The free end magnet can further reduce the resonant frequency of the ME antenna and improve the penetration ability of the antenna signal in the conductive medium. iii) Because permanent magnet vibrates itself under mechanical oscillation driven by AC electric field, it can be expected that vibrating magnet acts as another oscillation magnetic dipole besides the magnetostrictive dipole, which will result in the enhanced magnetic radiation.

Radiation Characterizations
Previous investigations have demonstrated that the radiation mechanism of the ME transmitter is based on the strain-induced magnetization oscillation and the electric polarization switching within the piezoelectric resonator. [10,33] Both the vibrating electric dipole and the magnetic dipole contribute to the electromagnetic radiation. To demonstrate the contribution of each component to overall radiation, near field radiation patterns were characterized. A circular loop receiver antenna was placed 20 cm away from the mechanical antenna in r, , and directions. For con-vention in the near-field radiation experiment, we orient the cantilevers with the longitudinal direction along the z-axis, the width direction along the x-axis, and the thickness direction along the y-axis (Figure 3e), respectively. For the ME cantilever, all the H r and H radiation come from the magnetic dipole. As the indicated in Figure 3b,c, the measured radiation patterns on the yz plane ( = 90°) and the xz plane ( = 0°) show the magnetic field H r reaching its maximum at = 0°and minimum at = 90°, while the magnetic field H reaches its maximum at = 90°a nd minimum at = 0°. The magnetic dipole does not have magnetic field radiation on the xy plane (Figure 3d), indicating that all the H component comes from the electric dipole. It is worth to note that the stress-free L-T mode ME composite can be regarded as ideal magnetic dipole, thus the radiation exhibits a symmetry pattern ( Figure S10, Supporting Information). However, due to the electromagnetic shielding effect of the aluminum alloy, the symmetry of the radiation pattern for ME cantilever is broken. It can be calculated from the B-field patterns that the directivity D of the ME cantilever is 3 dB. [5] The radiation signature of the MME antenna is significantly different than the ME cantilever. The additional oscillating magnetic dipole created by the physical motion of magnet at the free end of the ME antenna results in extra magnetic radiation. As shown in Figure 3f,g, the magnetic field radiation is significantly enhanced along the MME antenna's longitudinal direction close to the free end magnet, thus the space radiation symmetry of MME cantilever is further broken and the directivity is increased to 4 dB. That means the MME antenna tends to radiate in a certain direction. Moreover, the electric dipole induced radiation H remand unchanged, indicating the oscillating magnetic serves as pure magnetic dipole, and does not contribute to electric dipole radiation. Figure 4a describes the radiation field intensity for L-T mode ME composite, ME cantilever, and MME cantilever as a function of the power consumption, respectively. Because of the far-field region (kr>>1, and k is the wavenumber, r is the distance) at the resonant operating frequencies for the composites (24.6 kHz, 11.32 kHz, and 11.5 kHz, respectively) is beyond 2 km, it is hard to test the far-field radiation properties. Therefore, we investigated the near-field radiation to understand the radiation performance of our mechanical antenna. A search coil was located 20 cm away from the radiation source along the longitudinal direction, and the antennas were actuated with 500 mW at their resonance frequencies. The voltage output received from the search coil was divided by the transfer function of the coil to obtain the detected magnetic flux density. As expected, the radiation field intensity generated by the MME antenna is much stronger than the L-T mode ME composite and ME cantilever with the same applied power (Figure 4a).
It is worth noting that the rate of radiation field intensity increases as a function of the applied power gradually levels off, especially for the L-T mode ME composites. This is because the oscillation of the magnetic dipole in the magnetostrictive phase with electric induced strain could not increase all through. As a result, the magnetic radiation produced by the magnetostrictive phase of an ME-based antenna will eventually saturate. Oscillations of the magnet in the MME cantilever are not similarly limited and can thus produce larger oscillations as voltage further increased beyond the magnetostriction saturation power. As such, the MME cantilever antenna has the potential to generate high-power radiation than an ME-based device.
Then, the transmission capabilities of the MME cantilever have been investigated and shown in Figure 4b. In order to improve the sensitivity of the magnetic field receiving device, we used a ME magnetic field sensor to replace the search coil. The utilized ME sensor contains a sandwich structure ME composite, which consists of PZT piezoelectric fibers in series connected and two 4-layer Metglas films, as shown in Figure S11 (Supporting Information). The series connected mode can increase the electrical resistance by a large margin, thus leading to reduce the thermal noise. Figure S12 (Supporting Information) shows the electromechanical and ME coupling properties of the ME laminate. It can be seen low dielectric loss of 2% with small capacitance of ≈800 pF, which showing benefit in dielectric loss noise reduction. A pair of magnets were placed near the composite to provide bias magnetic field. The DME charge coefficients at Quasi-static and resonant states are measured to show the values of 1760 pC Oe −1 and 159 nC Oe −1 , respectively. Afterwards, a charge amplifier circuit was added to match the high impedance of the ME laminate. The whole ME sensor is illus-trated in Figure S13 (Supporting Information), which demonstrating a noise power spectrum density (NPSD) of 265 nV Hz −1/2 at the 11.32 kHz corresponding to the resonance frequency of MME cantilever ( Figure S14a, Supporting Information). With a 0.6 nT AC magnetic field (B AC ) excited at 11320 Hz, the signalto-noise-ratio (SNR) can be calculated to be 56 dB ( Figure S14c, Supporting Information). The limit of detection (LOD) has therefore been calculated as: LOD = B AC SNR = 0.6nT 56dB =0.95pT. Meanwhile, the magnetic field sensitivity (S) of the sensor is found to be 42 V Oe −1 at 11.32 kHz ( Figure S14b . Even though the sensor is not working at its resonance frequency, the ultralow equivalent magnetic noise is still enough to meet the requirements of receiving antenna signals. Figure S15 (Supporting Information) describes the actual test setup for the VLF communication test. The ME antenna was driven by a TREK Model 603 high-voltage power amplifier at 500 mW, while the ME sensor was connected to the Agilent 35670A signal analyzer. The magnetic field was measured by the ME sensor with varying distances of 0.1-5 m from the ME antenna. As shown in Figure 4b, magnetic flux rapid decreases with distance, while a ≈10 −11 T magnetic field can be detected at 5 m. Furthermore, the empirical curve fitting suggests that the measured magnetic flux decayed as 1/r 3 , which is well agreed with the decay law of quasi-static magnetic field. Previous work has shown that the radiation mechanism of the ME transmitter is mainly based upon the straininduced magnetization oscillation, and the decay law of magnetic flux also exhibits as quasi-static magnetic field. The B-field distribution of the MME cantilever, stress-free L-T mode ME laminate and one-end clamped ME cantilever under an identical driven power of 400 mW is theoretical compared ( Figure S16, Supporting Information). The comparison indicates the MME cantilever could generate much stronger magnetic flux at long distance, which directly proves the MME cantilever having superior performance over normal ME based antenna in long distance communication.
In order to understand the transmission capabilities, a theoretical current loop is used as comparison to get efficiency ratio of MME antenna. The loop area and input power are set to be the same as the area of MME antenna. As shown in Figure S17 (Supporting Information) empirical curve fitting suggests that the magnetic flux density produced by MME antenna and theoretical current loop both decayed as 1/r 3 , while the magnetic field produced by the loop (driven power of 500 mW, frequency at 11320 Hz) is over three orders smaller than the ME transmitter due to the non-resonance state and impedance mismatch. The efficiency of the MME antenna is estimated to be 4.8×10 −16 (see theoretical treatment in method). Additionally, the antenna gain G is calculated as G = D= − 149 dB. The current loop yields an efficiency ratio of 1.7×10 −23 and gain of −226 dB showing a drastically enhancement in antenna both efficiency and gain. Besides, the performance of the MME antenna in conductive medium of seawater was also theoretically demonstrated relative to the existing communication technology ( Figure S18a, Supporting Information). The attenuation of VLF MME antenna along the longitudinal direction in seawater is quicker than that in air (i.e., 1/r 3 , Figure S18b, Supporting Information). However, The MME antenna exhibits much better signal penetration ability due to the relatively low loss of VLF signal in conductive medium relative to the widely used 900 MHz and 5 GHz communication (see inset of Figure S18b, Supporting Information).

VLF Transmission
As the primary function of antenna is to realize effective wireless signal transmission, the signal modulation method for the new emerging MME antenna has become a non-negligible priority. First, we evaluated the ability to transmit analog signals directly using the MME antenna. Because of the continuity in the time domain, analog signal has infinite signal resolution theoretically, which means that analog signal has higher information density than digital signal. Meanwhile, Analog signal processing is simpler than digital signal processing, so that the transmission sys-tem does not need complex signal demodulation elements. To demonstrate analog signal transmission, we input an audio voltage signal to a power amplifier, and then the amplified signal is transmitted into the MME antenna (illustrated in Figure 5a). A coil is placed 15 cm from the cantilever to serve as a receiver. The received signal is amplified and fed to an oscilloscope for observation. When the MME antenna with fixed amplitude sine waves of various frequencies (0.5, 1, 3, and 5 kHz), the received signal exhibits a sinusoidal response of the same frequency (Figure 5b). Since the receiving coil produces larger voltage signal in response to high frequency magnetic field signal, we see an increase in amplitude of received signals with an increase in drive signal frequency. Similarly, an unmodulated, amplified audio signal input to the MME antenna yields a clear signal on the receiving coil that correlates highly with the drive signal ( Figure 5c) A loop antenna has also been tested as a comparison to the MME antenna and shows similar results. In Video S1 (Supporting Information), we demonstrate the wireless transmission of analog audio signals using the MME antenna and successfully played the received signals in a loudspeaker connected to the receiver coil. None of the signal transmission processes require signal modulation, indicating a portable strategy for analog signal near-field communication.
Another mode for signal transmission is digital modulation, which features protection against interference from other electronic sources and signals. [37,38] Simple and common digital modulation schemes can be utilized such as binary amplitude or frequency -shift keying (BASK, BFSK) to directly modulate the amplitude or frequency with a modulating bitstream. A mechanical antenna exhibits a response time under external acoustic actuation that is directly proportional to its quality factor and limits the ASK (on-off keying) rate since the mechanical system must be switched on and off corresponding to bit 1 and bit 0, respectively. This presents a tradeoff between the antenna quality factor Figure 6. Different digital signal modulation schemes using one-end clamped MME antenna. a) Schematic of the digital data transmission measurement setup. b) 10 Hz bit stream modulated by amplitude-shift keyed (ASK) method is input to the MME antenna and then received by a loop antenna followed by a demodulation process. c) Output voltage spectrum shows the received ASK signal consistent with the resonant carrier signal. d) 10 Hz bit stream modulated by frequency-shift keyed (FSK) method is input to the MME antenna and then received by a loop antenna followed by a demodulation process. e) Output voltage spectrum of the received FSK signal. and the maximum achievable data rate, which is required for efficient antenna radiation and bandwidth efficiency, respectively. To investigate the digital signal modulation application for MME antenna, ASK, and FSK modulation methods are tested as a final demonstration (Figure 6). Figure 6a shows a schematic of the digital data transmission measurement setup. The MME antenna is directly modulated using a function generator outputting both ASK and FSK signals with the resonant response of the MME antenna, a loop antenna is placed 15 cm away from the antenna and the received signals are demodulated using computer. For the ASK case shown in Figure 6b, a 10 Hz binary bit stream is converted to ASK modulation signal and loaded at the resonance frequency of the MME antenna. After demodulation and low-pass filtering, the measured signal is successfully converted to the original bit stream without distortion. Figure 6c shows the received voltage spectrum with the carrier signal at the resonance frequency of MME antenna. It is worth noting that as the driving signal is switched on and off the measured signal voltage ramps up and down over a duration inversely proportional to the loaded quality factor. The ramping time limits the fundamental modulation rate for ASK, thus leading to considerations in fundamental design tradeoff to balance the inversely proportional data rate with the high Q.
On the other hand, the FSK method can be designed to have a fixed amplitude and continuous phase, which mitigates the amplitude settling limitation compared to ASK modulation. Furthermore, anti-noise interference and anti-attenuation properties make FSK modulation more suitable for mechanical antennas modulation than ASK. As can be seen in Figure 6d, the FSK modulation was demonstrated by applying an FSK signal, which is converted from a 10 Hz binary bit stream, to the MME antenna. The carrier signals are 11.22 and 11.42 kHz, respectively, and the received voltage spectrum shown in Figure 6e clearly demonstrates the two carrier signal frequency peaks. In addition, an 11.22 kHz signal was also found because of the inherent oscillation from MME antenna induced by the switch between carrier signals. The original bit stream can be also recovered by the coherent demodulation process without bit error, indicating the compatibility with various modulation methods for the MME antenna.

Conclusion
In conclusion, we have demonstrated a magneto-mechanoelectric acoustically actuated antenna for efficient very low frequency (VLF) transmission. High quality factor PZT-4 www.advancedsciencenews.com www.advelectronicmat.de piezoelectric ceramic and multilayer FeSiB Metglas films are used for the ME composite preparation. Electric induced strain and ME coupling properties of ME composite with various Metglas layers are measured and the results indicate optimum ratio of magnetostrictive phase and piezoelectric phase enhanced ME coupling coefficient. After the ME composite was fixed to form a cantilever structure, a dramatic reduction in resonance frequency occurs compared to the L-T mode case, while the CME coupling is compromised because of the dispersion of vibrational energy. Two magnets were utilized in the MME cantilever fabrication. It was found the attached magnets not only provide optimum bias magnetic field and further reduce the resonance frequency to 11.32 kHz, but also significantly enhanced the ME cantilever radiation capability through the contribution of physical magnet oscillation. The MME antenna shows a dramatic increase in efficiency and gain as compared to a simple loop antenna, particularly at resonant frequencies. Analog signal transmission without modulation utilizing MME antenna was demonstrated, which suggests a convenient strategy for analog signal near field communication. Finally, ASK and FSK digital signal modulation were conducted with an 11.3 kHz carrier signal to demonstrate anti-interference and anti-attenuation communication modulation capabilities.

Experimental Section
Materials Preparation: In this work, a commercially purchased PZT-4 piezoelectric ceramic and amorphous FeSiB alloys (Metglas) (Vacuumschmelze GmbH & Co. KG, Germany) were chosen as the piezoelectric phase and magnetostrictive phase (respectively) for the magnetoelectric composite fabrication. The PZT-4 ceramic was first poled with a high-voltage power supply (MPD-10 kV, Partulab Technology Co.Ltd). The 0.022 mm thick Metglas films were cut to a length of 70 mm and a width of 6 mm As-cut Metglas layers were multi-stacked using epoxy resin (West system 206, USA) and then cured for 24 h at room temperature to form the 5-20 multi-layer Metglas films. Then the PZT was bonded on the Metglas beam with epoxy resin (West system 206, USA) and cured at room temperature for 24 h under even pressure. For the MME cantilever fabrication, the L-T mode ME composite was fixed to a 10 cm by 8 cm aluminum plate using epoxy resin. A magnet was attached to the free end of the beam and to the aluminum plate to provide bias magnetic field with parameters determined from FEA simulation. The geometry parameter of the mechanical antenna is illustrated in Figure 1a. An ME magnetic field sensor was used for near-field magnetic radiation detection. PZT-5 ceramic fibers with size of 40 mm×2 mm×0.5 mm were chosen for the piezoelectric phase fabrication. 7 PZT fibers were connected in series with flexible printed circuit to form a piezoelectric macrofiber composite (MFC). 2 pieces of 4-layer Metglas films were attached to the MFC using epoxy resin (West system 206, USA) and cured at room temperature for 24 h under even pressure. A charge amplifier circuit was connected to the manufactured ME composite and two bias magnets were placed near the ME composite to form the final sensor.
Characterization of ME Performance: For the electric field-induced strain measurement, a macrostrain gauge was attached on the surface of PZT and ME laminates using adhesive. The strain value was collected using dynamic strain meter (AFT-0951, China). The DME coefficient of the ME composite as a function of DC bias field (H DC ) was measured using lock-in amplifier (SR830, Stanford Research Systems, USA) under fixed AC magnetic field (H AC ). The H DC was supplied using a Helmholtz coil powered by a DC current source (KEPCO, China), and the H AC was provided by a solenoid driven via lock-in amplifier. The DME coefficient measurement setup is shown in Figure S9 (Supporting Information).

Antenna Radiation Measurement:
In this part, the as made ME magnetic field sensor was used as the long-distance receiver to capture magnetic radiation signals. The ME antenna and MME antenna were powered by a power amplifier (ATA-3090, Aigtek) and a signal generator (DG1022, Rigol). The measured radiation signals were characterized using an Agilent 35670A signal analyzer.
Analog and Digital Signal Transmission: An analog audio signal without modulation was amplified by a power amplifier (ATA-3090, Aigtek) and input to MME antenna. A 200-turn coil with a diameter of 10 mm and length of 30 mm was used as the receiver. The signal measured by the receiver coil was amplified and input to a loudspeaker to demonstrate wireless audio transmission. For digital signal transmission, a binary bit stream was converted to ASK or FSK signal using a function generator (DG1022, Rigol), and then amplified by a power amplifier (HA-805, Pintech). The magnetic radiation signal was measured using a loop antenna and demodulated by computer.
Calculation of Converse Magnetoelectric Coefficient: Stress-free L-T mode composite, ME cantilever, and MME cantilever are considered in this paper. For the L-T mode ME composite, the material equations for the piezoelectric, magnetostrictive, and passive phase were built to be: , mT 1 = m Y( m S 1 − q 11 h 1 ), m B 1 = 0 h 1 + q m 11 T 1 , and s T 1 = s S 1 s s 11 , respectively. Where p T 1 , s T 1 , and m T 1 are the stress in the piezoelectric, passive, and magnetostrictive phase, p S 1 and s S 1 are the strain for the piezoelectric and passive phase, p s 11 and s s 11 are the elastic compliance constant for the piezoelectric and passive phase, d 31 is the transverse piezoelectric coefficient, E 3 is the applied electric field, q 11 is the piezomagnetic coefficient, h 1 is the variable magnetic field strength arising during the inverse magnetoelectric effect in the magnetostrictive phase, m B 1 is variable magnetic induction in the magnetoelectric structure, m Y is the Young's modulus of magnetostrictive phase, and μμ 0 is permeability. , L is the length of ME composite).
For the ME cantilever, the material equations and motion equation are same with L-T mode composite, while the boundary conditions are U x (0) = 0, T 1 (L) = 0. The converse ME coefficient for ME cantilever is .
For the MME cantilever, the calculation was divided into two parts (ME composite and magnet). For the first part, the material equations and motion equation are same with L-T mode composite and ME cantilever.
For the second part (magnet), the motion equation is a 2 U x2 t 2 = a T 12 x , − a 2 U x2 = a Y 2 U x2 x 2 ( a is the magnet density, U x2 longitudinal displacement of magnet, a T 12 is the stress tensor of the second part). For the joints of the first and second sections, the following boundary conditions are valid: U x1 (0) = U x2 (0),F 11 (0) = F 12 (0), and longitudinal force on the first and second part are: F 11 (0) = T 11 (0)t 1 w, F 12 (0) = a T 12 (0) D 2 4 , where T 11 is the stress tensor of the first part, w is the width of the ME composite, D is the permanent magnet diameter. Finally, the converse ME coefficient for the MME cantilever is (More details can be found in Section S1, Supporting Information): Calculated Antenna Efficiency, Directivity, and Gain: Here a small circular loop antenna was used for comparison to obtain the efficiency of MME antenna. The near field magnetic flux density of the small circular loop